Method for digital control of a universal motor, in particular for electrical household appliances

ABSTRACT

The invention concerns a method for digital control of a universal motor, in particular for electrical household appliance, comprising steps which consist on: measuring said engine ( 21 ) rotation speed; determining the difference between the measured speed and a set speed; and controlling the motor on the basis of said difference. Said method further comprises a step ( 24, 25 ) which consists in estimating at least one of the values of the resisting torque (C) and the current (i) in the motor windings. The invention is useful for controlling washing machines.

The present invention relates to a method for the digital control of auniversal motor, notably for domestic use, and more particularly such amethod comprising the steps consisting in

measuring the speed of rotation of the said motor,

determining the difference between the measured speed and a referencespeed, and

controlling the said motor according to the said difference.

Universal motors are single-phase motors with a series excitationcommutator, and are frequently used in the field of domestic electricalequipment, notably for driving the drums of washing or drying machines.

The regulation of the speed of rotation of a universal motor involves atachometric generator coupled to this motor in order to supply as anoutput an electrical signal whose frequency is proportional to thespeed, and a supply triac whose triggering angle determines the meanvoltage applied to the motor windings. A control law makes it possibleto adapt the triggering angle according to the speed.

Digital control methods have already been proposed. Currently, the lawused is very simple since a triggering angle difference proportional tothe speed difference is adopted:

α_(n+1)=α_(n) +A·Δω  (1)

where

α_(i) is the triggering angle of the triac at time t_(i)

Δω is the difference between the speed measured by the tachometricgenerator and the reference speed, and

A is a proportionality coefficient.

In addition, provision is generally made for limiting the current in themotor. To this end, a shunt makes it possible to measure this current.

Although generally giving satisfaction, this method has a certain numberof drawbacks.

Firstly, it provides no access to the physical quantities other thanthose which are directly measured, namely essentially the speed ofrotation and possibly the current in the motor. The known method istherefore relatively incomplete.

In addition, knowledge of the current requires, in addition to thepresence of the shunt, knowledge of an amplifier and current to voltageconverter. The result is a complication of the circuit and consequentlyan increase in costs. In particular, it cannot supply the true effectivecurrent without fairly powerful calculation means.

However, this knowledge of the current is particularly useful in certainapplications, independently of the need to limit it, and notably in thecase of the driving of the drum of a washing machine. This is because,in the case of a universal motor, the torque is directly a function ofthis current. And knowing the torque gives access to the load, on theassumption that the motor follows the acceleration inputs.

The load is a quantity, knowledge of which makes it possible to regulatethe speed in an optimum fashion in the aforementioned case of thedriving of the drum of a washing machine. This is because theeffectiveness of the washing is to a major extent a function of therelative speeds of the washing and the water contained in the drum.However, when the load is low, the washing falls by itself in thewashing water during its rotation. The control law can thereforeadvantageously be adapted to the machine load.

The present invention aims to mitigate the drawbacks set out above.

More particularly, the invention aims to provide a digital controlmethod for a universal motor, notably for domestic use, which dispenseswith the use of the shunt of the prior art.

The invention also aims to provide such a control method which makes itpossible to take account of the motor load without requiring any currentmeasurement.

In addition, the invention aims to provide such a control method whichmakes it possible, in the case of a washing or drying machine, to takeaccount of the imbalance created by an irregular arrangement of thewashing.

To this end, the object of the invention is a method for the digitalcontrol of a universal motor, notably for domestic use, comprising thesteps consisting in

measuring the speed of rotation of the said motor,

determining the difference between the measured speed and a referencespeed, and

controlling the said motor according to the said difference,

characterised by the fact that it also comprises a step of estimating atleast one of the quantities including the resistive torque (C) and thecurrent (i) in the motor windings.

It will be seen that it is possible to deduce the motor load from theresistive torque (C), which is particularly advantageous in the case ofa domestic electrical appliance in which this parameter is generallyunknown to the user.

The method according to the invention can also comprise the estimationof at least the moment of inertia or the coefficient of viscousfriction.

It is also possible to make provision for the estimation of theresistive torque to comprise the estimation of the sum of asubstantially constant mean torque and a sinusoidal imbalance component.

It is then possible, in the case of a washer, to take account of anirregular arrangement of the washing in the drum in order to obtain amore even speed of rotation of the latter.

Also in a particular embodiment, the method of the invention cancomprise the step consisting in determining an estimation, possiblyoptimum, of the speed of rotation of the motor.

Also in a particular embodiment, it may comprise the step consisting indetermining an estimation of the motor load.

Such estimations make it possible in particular to provide an optimumcontrol taking account also of the load and operating temperature of themotor.

In particular, the method according to the invention can comprise thestep consisting in determining the cycle ratio of the motor operatingcycle. Cycle ratio of the motor operating cycle means here the ratiobetween the actual operating time and the total time of the cycle,including the time during which the motor is stopped.

In a first embodiment of the invention, the motor speed regulation iseffected by any known means. It is possible in particular to determinethe triggering angle of the triac by a formula such as formula (1).

It is possible on the other hand, in another embodiment of theinvention, to include the aforementioned estimations in the regulationloop. The control voltage to be applied to the motor is then calculatedaccording to these estimations, and the triggering angle of the triac isderived from this calculated voltage.

Consequently, in the method according to this embodiment of theinvention, there is determined, not directly and empirically thetriggering angle of the triac, but the voltage to be applied to themotor input. Being in possession of this voltage, it is possible toderive therefrom, in addition to the triggering angle, variouselectrical and mechanical operating parameters, and thus to optimisecontrol.

A description will now be given, by way of non-limitative examples, ofparticular embodiments of the invention, with reference to theaccompanying schematic drawings, in which:

FIG. 1 is a diagram of a washing machine provided with a control deviceimplementing the method according to the invention;

FIG. 2 is a block diagram of the control; and

FIG. 3 is a flow diagram of the method.

The first step will be to describe the modelling of the device, and thena control law according to the invention, and finally various types ofparameter estimators. Next a description will be given of the practicalimplementation of the invention, taking account in particular of thedigital character of the control.

Modelling

There can be seen in FIG. 1 a washing machine comprising a drum 1disposed in a washing tank (not shown) and driven by a universal motor2. A tachometric generator 3 is mounted on the shaft of the motor 2 inorder to supply, to a control circuit 4, an electrical signal with afrequency proportional to the speed of rotation ω of the motor.

The motor 2 is supplied from the mains by means of a triac 5. Thecircuit 4 controls the triac 5 after having determined the requiredtriggering angle α as a function of the speed ω and a reference speedω_(c). To this end, the circuit 4 includes the necessary circuits andnotably a microcontroller.

The universal motor 2 can be modelled by the system of the following twodifferential equations, translating respectively the mechanical andelectrical functioning of the motor $\left\{ \begin{matrix}{{{{J} \cdot \overset{.}{\omega}} + {F \cdot \omega}} = {{k_{1} \cdot i^{2}} - C}} & \quad & (2) \\{{{L\frac{i}{t}} + {Ri} + {k_{2}{i \cdot \omega}}} = v} & \quad & (3)\end{matrix} \right.$

where

{dot over (ω)} represents the derivative of ω with respect to the time$\frac{\omega}{t},$

J represents all the inertias related to the motor,

F is the coefficient of viscous friction,

C is the resistive torque comprising all the torques not otherwisemodelled, notably the solid frictions and the fluctuations in torque dueto the movement of the washing in the drum and any imbalance caused byan irregular arrangement of the washing,

L and R are respectively the inductance and resistance of the motor

i and J are respectively the current passing through the motor and thevoltage at its terminals, and

k₁ and k₂ are constant coefficients.

If simplifying assumptions are made according to which the moment ofinertia J is constant and therefore does not depend on the weight ofwashing present in the drum,

the frictions are purely viscous frictions proportional to the speed,

the term relating to the inductance is negligible compared with theother terms of the second equation, and

the coefficients k₁ and k₂ are substantially equal within the limits ofuse of the motor, which is proved by the fact that they are related tothe geometry of the motor,

the following simplified equations are obtained $\left\{ \begin{matrix}{{{J \cdot \overset{.}{\omega}} + {F \cdot \omega}} = {{k \cdot i^{2}} - C}} & \quad & (4) \\{{{Ri} + {k \cdot \omega \cdot i}} = v} & ~ & (5)\end{matrix} \right.$

where F is a constant.

Control Law

From the above two equations (4) and (5), it is deduced that$\begin{matrix}{\overset{.}{\omega} = {{{- \frac{F}{J}}\omega} + {\frac{k}{J}\frac{v^{2}}{\left( {R + {k \cdot \omega}} \right)^{2}}} - {\frac{1}{J}C}}} & (6)\end{matrix}$

In addition, it is wished for the control law to make the behaviour ofthe system linear, that is to say that it is such that

{dot over (ω)}=−k.(ω−ω_(c))  (7)

where K is a coefficient to be chosen by experimentation. In otherwords, K is the inverse of the time constant for the reduction in thedifference between the measured speed and the reference speed ω_(c), Thevalue of K is chosen so as to obtain from the system the requiredbehaviour in response to the given command. The coefficient K can alsovary as a function of ω_(c).

Thus there is obtained $\begin{matrix}{v = {\sqrt{\frac{1}{k}\left\lbrack {{{- K} \cdot J \cdot \left( {\omega - \omega_{c}} \right)} + C + {F \cdot \omega}} \right\rbrack}*\left( {R + {k \cdot \omega}} \right)}} & (8)\end{matrix}$

which is the control law giving the voltage J to be applied to the motoras a function of the measured speed of rotation ω, the reference speedω_(c) and the resistive torque C.

It is possible to introduce in addition an integral effect by adding thestate y such that

{dot over (y)}=ω−ω _(c)  (9)

which leads to the control law $\begin{matrix}{v = {\sqrt{\frac{1}{k}\left\lbrack {{{- K} \cdot J \cdot \left( {\omega - \omega_{c}} \right)} + C + {F \cdot \omega} - {K_{1} \cdot y}} \right\rbrack}*\left( {R + {k \cdot \omega}} \right)}} & (10)\end{matrix}$

Naturally, this equation can be used only if

−K.J.(ω−ω_(c))+C+F.ω−K ₁ .y≧0  (11)

which occurs only for values of the speed of rotation greater than acertain limit, a function itself, moreover, of the speed difference andits derivative.

In the contrary case, $\begin{matrix}{v = {\sqrt{\frac{1}{k}\left\lbrack {C + {F \cdot \omega} - {K_{1} \cdot y}} \right\rbrack}*\left( {R + {k \cdot \omega}} \right)}} & (12)\end{matrix}$

will be adopted for the control law.

The latter is itself valid only if

C+F.ω−K ₁ .y≧0  (13)

Finally, if the last equality is not satisfied, will be taken:$\begin{matrix}{v = {\sqrt{\frac{1}{k}\left\lbrack {{F \cdot \omega} - {K_{1} \cdot y}} \right\rbrack}*\left( {R + {k \cdot \omega}} \right)}} & (14)\end{matrix}$

It will be seen below that it is not essential to assume that J and Fare constants, subject to the introduction into the description of thesystem of the corresponding supplementary states.

Parameter Estimators

Whatever the control law adopted, it is necessary to obtain anestimation C of the resistive torque, that is to say to produce anestimator of the torque. More generally, it would be possible to seek toestimate the values of all the parameters not assumed to be constant.

Two approaches for obtaining such an estimator will be described. Thefirst approach consists in minimising an energy related to thedifference between the real values of the parameters and their estimatedvalue. The other approach amounts to applying the state equations to theestimated values of the parameters.

First Approximation

First of all take the context of the above assumption, where the momentof inertia J and the coefficient of friction F are assumed to beconstant and known. An integral estimator is produced, in the form

{circumflex over ({dot over (C)})} ₀ =−K′.J.(ω−ω_(c))  (15)

In other words, the estimation is readjusted as a function of theobserved difference between the measured speed and the reference speed.

The value of K′ is chosen so that the estimator behaves in the requiredmanner.

If on the other hand it is now assumed that the moment of inertia J andthe coefficient of viscous friction F are no longer constants, they arereplaced by their estimations, respectively Ĵ and {circumflex over (F)}.

We then put $\begin{matrix}{{{\hat{J} \cdot \overset{.}{\omega}} + {\hat{F} \cdot \omega}} = {\frac{k \cdot v^{2}}{\left( {R + {k \cdot \omega}} \right)^{2}} - \hat{C}}} & (16)\end{matrix}$

which gives $\begin{matrix}{{{\hat{J} \cdot {\overset{.}{\omega}}_{c}} + {\hat{F} \cdot \omega_{c}}} = {\frac{k \cdot v^{2}}{\left( {R + {k \cdot \omega}} \right)^{2}} - \hat{C} + {K \cdot \overset{\sim}{\omega}}}} & (17)\end{matrix}$

where {tilde over (ω)} is the error on the estimation {circumflex over(ω)}: $\begin{matrix}{\overset{\sim}{\omega} = {\hat{\omega} - \omega}} & (18)\end{matrix}$

The general control law is thus obtained $\begin{matrix}{v = {\sqrt{\frac{{\hat{F} \cdot \omega_{c}} + \hat{C} - {K \cdot \overset{\sim}{\omega}} + {\hat{J} \cdot {\overset{.}{\omega}}_{c}}}{k}}*\left( {R + {k \cdot \omega}} \right)}} & (19)\end{matrix}$

As before, this control law is valid only if $\begin{matrix}{\frac{{\hat{F} \cdot \omega_{c}} + \hat{C} - {K \cdot \overset{\sim}{\omega}} + {\hat{J} \cdot {\overset{.}{\omega}}_{c}}}{k} \geq 0} & (20)\end{matrix}$

If this test is not satisfied, then the same control laws as above areadopted.

There are then taken, in order to estimate the different terms$\left\{ \begin{matrix}{\overset{.}{\hat{C}} = {{- \frac{1}{\tau_{1}}}\quad \overset{\sim}{\omega}}} & {\quad (21)} \\{\overset{.}{\hat{F}} = {{- \frac{1}{\tau_{2}}}\quad {\omega_{c} \cdot \overset{\sim}{\omega}}}} & {\quad (22)} \\{\overset{.}{\hat{J}} = {{- \frac{1}{\tau_{3}}}{\left( \overset{.}{\omega} \right)_{c} \cdot \overset{\sim}{\omega}}}} & {\quad (23)\quad}\end{matrix} \right.$

Second Approximation

It will be assumed firstly that the mean torque C₀ is constant incontinuous running and consequently that {dot over (C)}₀=0. If C isconsidered to be a supplementary state, there is written$\left\{ \begin{matrix}{\overset{.}{\omega} = {{{- \frac{F}{J}}\quad \omega} - {\frac{1}{J}\quad C} + {\frac{k}{J}\quad \frac{v^{2}}{\left( {R + {k \cdot \omega}} \right)^{2}}}}} & {\quad (24)} \\{\overset{.}{C} = 0} & {\quad (25)}\end{matrix} \right.$

The estimator of the angular speed {circumflex over (ω)} and of theresistive torque Ĉ are defined by the following equations$\left\{ \begin{matrix}{\overset{.}{\hat{\omega}} = {{{- \frac{F}{J}}\quad \hat{\omega}} - {\frac{1}{J}\quad \hat{C}} + {\frac{k}{J}\quad \frac{v^{2}}{\left( {R + {k \cdot \omega}} \right)^{2}}} + {L_{1}\left( {\omega - \hat{\omega}} \right)}}} & {\quad (26)} \\{\overset{.}{\hat{C}} = {L_{2}\left( {\omega - \hat{\omega}} \right)}} & {\quad (27)}\end{matrix} \right.$

which amounts to estimating both ω and C using the differenceinformation {tilde over (ω)}.

If {tilde over (ω)} and {tilde over (C)} are the errors on theseestimations, equations (24) to (27) are derived $\left\{ \begin{matrix}{\overset{.}{\overset{\sim}{\omega}} = {{{- \frac{F}{J}}\quad \overset{\sim}{\omega}} - {\frac{1}{J}\quad \overset{\sim}{C}} - {L_{1}\left( {\omega - \hat{\omega}} \right)}}} & {\quad (28)} \\{\overset{.}{\overset{\sim}{C}} = {- {L_{2}\left( {\omega - \hat{\omega}} \right)}}} & {\quad (29)}\end{matrix} \right.$

where L₁ and L₂ are the gains of the estimator.

A matrix Λ is derived from this, such that $\begin{matrix}{\begin{pmatrix}\overset{.}{\overset{\sim}{\omega}} \\\overset{.}{\overset{\sim}{C}}\end{pmatrix} = {\Lambda \cdot \begin{pmatrix}\overset{\sim}{\omega} \\\overset{\sim}{C}\end{pmatrix}}} & (30)\end{matrix}$

where the matrix Λ is defined by $\begin{matrix}{\Lambda = \begin{pmatrix}{{- \frac{F}{J}} - L_{1}} & {- \frac{1}{J}} \\{- L_{2}} & 0\end{pmatrix}} & (31)\end{matrix}$

Let P be the polynomial characteristic of the estimator $\begin{matrix}{P = {{\det \left\lbrack {{\lambda \cdot I} - \Lambda} \right\rbrack} = {{\lambda^{2} - {\lambda \cdot \left( {{- \frac{F}{J}} - L_{1}} \right)} - \frac{L_{2}}{J}} = {\left( {\lambda - p_{1}} \right) \cdot \left( {\lambda - p_{2}} \right)}}}} & (32)\end{matrix}$

p₁ and p₂ being the poles of the error of the estimator which mustensure the convergence towards zero of the estimation error. These polesare chosen so as to obtain the best estimation and the gains L₁ and L₂are derived therefrom by $\left\{ \begin{matrix}{L_{1} = {{- \left( {p_{1} + p_{2}} \right)} - \frac{F}{J}}} & {\quad (33)} \\{L_{2} = {{- p_{1}} \cdot p_{2} \cdot J}} & {\quad (34)}\end{matrix} \right.$

It will now be considered, on the other hand, that the resistive torqueis the sum of a mean torque {overscore (C)}₀ which is constant incontinuous running and an oscillator τ whose amplitude and phasedifference will be determined so that the command opposes the imbalancecreated by the mass of wet washing. It will be assumed for this purposethat the imbalance is a sinusoidal signal whose angular frequency isequal to the mean angular frequency of the drum.

We then put

Ĉ={overscore (C)} ₀ +z  (35)

with

 {umlaut over (z)}ω ² .z=0  (36)

where

{dot over (z)}=z ₁  (37)

{dot over (z)} ₁=−ω² .z  (38)

If the three supplementary states C, z and z₁ there is written$\left\{ \begin{matrix}{\overset{.}{\omega} = {{{- \frac{F}{J}}\quad \omega} - {\frac{1}{J}\quad C_{0}} - {\frac{1}{J}\quad z} + {\frac{k}{J}\quad \frac{v^{2}}{\left( {R + {k \cdot \omega}} \right)^{2}}}}} & {\quad (39)} \\{\overset{.}{C} = 0} & {\quad (40)} \\{\overset{.}{z} = z_{1}} & {\quad (41)} \\{{\overset{.}{z}}_{1} = {{- \omega^{2}} \cdot z}} & {\quad (42)}\end{matrix} \right.$

It is therefore possible to write the estimator $\left\{ \begin{matrix}{\overset{.}{\hat{\omega}} = {{{- \frac{F}{J}}\quad \hat{\omega}} - {\frac{1}{J}\quad {\hat{C}}_{0}} - {\frac{1}{J}\quad \hat{z}} + {\frac{k}{J}\quad {\hat{i}}^{2}} + {L_{1}\left( {{\omega--}\hat{\omega}} \right)}}} & {\quad (43)} \\{{\overset{.}{\hat{C}}}_{0} = {L_{2}\left( {{\omega--}\hat{\omega}} \right)}} & {\quad (44)} \\{\overset{.}{\hat{z}} = {{\hat{z}}_{1} + {L_{3}\left( {{\omega--}\hat{\omega}} \right)}}} & {\quad (45)} \\{{\overset{.}{z}}_{1} = {{{- \omega^{2}} \cdot \hat{z}} + {L_{4}\left( {{\omega--}\hat{\omega}} \right)}}} & {\quad (46)}\end{matrix} \right.$

As before, the errors on the estimations are written in the form$\begin{matrix}{\begin{pmatrix}\overset{.}{\overset{\sim}{\omega}} \\\overset{.}{\overset{\sim}{C}} \\\overset{.}{\overset{\sim}{z}} \\{\overset{.}{\overset{\sim}{z}}}_{1}\end{pmatrix} = {\Lambda \cdot \begin{pmatrix}\overset{\sim}{\omega} \\\overset{\sim}{C} \\\overset{\sim}{z} \\{\overset{\sim}{z}}_{1}\end{pmatrix}}} & (47)\end{matrix}$

with $\begin{matrix}{\Lambda = \left( \quad \begin{matrix}{{- \frac{F}{J}} - L_{1}} & {- \frac{1}{J}} & {- \frac{1}{J}} & 0 \\{- L_{2}} & 0 & 0 & 0 \\{- L_{3}} & 0 & 0 & 1 \\{- L_{4}} & 0 & {- \omega^{2}} & 0\end{matrix}\quad \right)} & (48)\end{matrix}$

The characteristic polynomial of the matrix Λ is $\begin{matrix}\begin{matrix}{P = \quad {{\det\left\lbrack {{\lambda \cdot I} - \Lambda} \right\rbrack} = {\lambda^{4} + {\frac{{L_{1} \cdot J} + F}{J}\quad \lambda^{3}} + {\frac{{- L_{3}} - L_{2} + {J \cdot \omega^{2}}}{J}\quad \lambda^{2}} +}}} \\{\quad {{\frac{{L_{1} \cdot J \cdot \omega^{2}} - L_{4} + {F \cdot \omega^{2}}}{J}\quad \lambda} + \frac{{- L_{2}} \cdot \omega^{2}}{J}}} \\{= \quad {\left( {\lambda - p_{1}} \right) \cdot \left( {\lambda - p_{2}} \right) \cdot \left( {\lambda - p_{3}} \right) \cdot \left( {\lambda - p_{4}} \right)}}\end{matrix} & (49)\end{matrix}$

As before the poles p₁ to p₄ of the error of the estimator are chosen asa function of the required response and the gains are deduced therefromby $\left\{ \quad {\begin{matrix}{L_{1} = {p_{1} - \frac{F}{J}}} & (50) \\{L_{2} = {{- \frac{J}{\omega^{2}}}\quad p_{4}}} & (51) \\{L_{3} = {{{- J} \cdot p_{2}} + {\frac{J}{\omega^{2}}\quad p_{4}} + {J \cdot \omega^{2}}}} & (52) \\{{L_{4} = {{J \cdot \omega^{2} \cdot p_{1}} - {J \cdot p_{3}} + {F \cdot \omega^{2}} - {{J \cdot \omega}\quad \frac{F}{J}}}}\quad} & (53)\end{matrix}\quad } \right.$

Implementation

A description will now be given, with reference to the block diagram ofFIG. 2 and the flow diagram in FIG. 3, of a particular implementation ofthe invention.

There can be seen in FIG. 2 a sampler 6 which samples the values of thespeed ω at intervals of time Δ_(t). From these speed samples thesuccessive values of the speed reference ω_(c) are deducted, in asubtracter 7. The speed difference ω−ω_(c) is applied to the input ofthe unit 8 establishing the control law, which determines the voltage vto be applied to the motor 2. A voltage limiter 9 is however disposeddownstream of the unit 8 in order to limit the voltage to valuesacceptable to the motor 2.

A block 10 determines, from the value of the voltage, possibly limited,the triggering angle α of the triac 5.

Taking account of its mechanical and electrical characteristics, thesystem 11, comprising essentially the washer and its content, adopts thespeed ω.

The sampled speed is also used in the estimator 12 in order to supplythe estimated torque Ĉ. This estimation is itself supplied to the unit 8for calculating the voltage to be applied to the motor 2.

It will be observed that, as mentioned previously, the estimator 12could also estimate the coefficient of friction F and the moment ofinertia J if the latter were not considered to be constant. Theestimated values {circumflex over (F)} and Ĵ would then in this casealso be supplied to the calculation unit 8.

It will also be observed that the subtracter 7 receives the measuredspeed ω directly as an input. It would however be possible to conceiveestimating this speed with an optimum estimator such as a Kalman filter.

Reference will now be made to FIG. 3.

Step 20 of the method consists in initialising the different values ofthe parameters when the device is started up. This step is obviously notreiterated at the following cycles.

Each cycle, of a duration Δt equal to the sampling period, commenceswith the capture of the speed of rotation {overscore (ω)} at 21 and thecalculation of the voltage v_(t) to be applied to the motor at 22. Thiscalculation is made for example using equations (10) to (14), replacingthe speed ω in the equations with the measured speed ω_(t) and thetorque C with the torque Ĉ estimated at the previous cycle.

From v_(t), the triggering angle α_(t) is determined at 23, by means ofa formula of the type

α_(t) =F ⁻¹(v _(t))  (54)

which can for example define a linear equation, whose coefficients willbe determined by experimentation.

Next the current is estimated at 24 by the equation $\begin{matrix}{{\hat{i}}_{1} = \frac{v_{t}}{R + {k \cdot \omega_{t}}}} & (55)\end{matrix}$

and then at 25 the torque, by means of the following equations derivedfrom equations (26) and (27) $\left\{ \quad {\begin{matrix}{{\hat{\omega}}_{t + 1} = {{\left\lbrack {1 - {\Delta_{T}\left( {\frac{F}{J} + L_{2}} \right)}} \right\rbrack \cdot {\hat{\omega}}_{t}} - {\frac{\Delta_{T}}{J}\quad {\hat{C}}_{t}} + {\frac{k \cdot \Delta_{T}}{J}\quad {\hat{i}}^{2}} + {\Delta_{T} \cdot L_{1} \cdot \omega_{t}}}} & (56) \\{{\hat{C}}_{t + 1} = {{\hat{C}}_{t} + {\Delta_{T} \cdot {L_{2}\left( {\omega_{t} - {\hat{\omega}}_{t}} \right)}}}} & (57)\end{matrix}\quad } \right.$

where L₁ and L₂ are determined by equations (33) and (34).

At 26, the newly calculated estimations {circumflex over (ω)}_(t+1) andĈ_(t+1) are stored in place of {circumflex over (ω)}_(t) and Ĉ_(t) and anew cycle can commence.

By virtue of the estimation of the intensity i, it is possible to limitthe current in the windings of the motor. If in fact the estimationexceeds a critical value, the triggering angle α is modified so as tobring the current down to an acceptable value. It will be noted that itis thus possible to control the current without measuring it, which wasone of the objectives of the invention. Moreover, account has not beentaken here of the factor Ldi/dt. Such a taking into account wouldhowever not present any particular difficulty if the precision of theestimation so required.

Having estimated values for the current and for the torque has manyadvantages.

With regard to the current, it makes it possible to determine thedissipated energy R.i² and consequently also to estimate the temperatureof the motor. Account can be taken of this in order to determine thecycle ratio of the operating cycle of the motor.

The estimation of the torque also supplies an estimation of the load onthe machine. There too, it is possible to use this information in orderto determine the aforementioned cycle ratio. An optimisation of thiscyclic ratio can moreover be sought as a function of both thetemperature and load estimations.

What is claimed is:
 1. A method for the digital control of a universalmotor, having a triac, the method comprising the steps of: measuring thespeed of rotation (ω) of the said motor, determining the differencebetween the measured speed and a reference speed (ω_(c)), andcontrolling a triggering angle of the triac as a function of saiddifference, characterized by the fact that the step of determining thedifference between the measured speed and the reference speed comprisesa step of estimating at least one of the quantities comprising theresistive torque (C) and the current (i) in the motor windings.
 2. Amethod for the digital control of a universal motor according to claim1, in which the determination (22) of a voltage to be applied to themotor also comprises the estimation of at least the moment of inertia(J) of the coefficient of viscous friction (F).
 3. A method for thedigital control of a universal motor according to claim 1, in which theestimation on the resistive torque (C) comprises the estimation of thesum of a substantially constant mean torque and an imbalance oscillator.4. A method for the digital control of a universal motor according toclaim 2, in which the estimation on the resistive torque (C) comprisesthe estimation of the sum of a substantially constant mean torque and animbalance oscillator.
 5. A method for the digital control of a universalmotor according to claim 1, comprising the step consisting indetermining an estimation of the speed of rotation of the motor.
 6. Amethod for the digital control of a universal motor according to claim2, comprising the step consisting in determining an estimation of thespeed of rotation of the motor.
 7. A method for the digital control of auniversal motor according to claim 3, comprising the step consisting indetermining an estimation of the speed of rotation of the motor.
 8. Amethod for the digital control of a universal motor according to claim1, comprising the step consisting in determining an estimation of theload on the motor.
 9. A method for the digital control of a universalmotor according to claim 2, comprising the step consisting indetermining an estimation of the load on the motor.
 10. A method for thedigital control of a universal motor according to claim 3, comprisingthe step consisting in determining an estimation of the load on themotor.
 11. A method for the digital control of a universal motoraccording to claim 5, comprising the step consisting in determining anestimation of the load on the motor.
 12. A method for the digitalcontrol of a universal motor according to claim 1, comprising the stepconsisting in determining the cycle ratio of the motor operating cycle.13. A method for the digital control of a universal motor according toclaim 2, comprising the step consisting in determining the cycle ratioof the motor operating cycle.
 14. A method for the digital control of auniversal motor according to claim 3, comprising the step consisting indetermining the cycle ratio of the motor operating cycle.
 15. A methodfor the digital control of a universal motor according to claim 5,comprising the step consisting in determining the cycle ratio of themotor operating cycle.
 16. A method for the digital control of auniversal motor according to claim 8, comprising the step consisting indetermining the cycle ratio of the motor operating cycle.
 17. A methodfor the digital control of a universal motor according to claim 1,comprising the steps consisting in: calculating, according to saidestimations, the control voltage to be applied to the motor, andderiving the triggering angle of the triac from this calculated voltage.18. A method for the digital control of a universal motor according toclaim 2, comprising the steps consisting in: calculating, according tosaid estimations, the control voltage to be applied to the motor, andderiving the triggering angle of the triac from this calculated voltage.19. A method for the digital control of a universal motor according toclaim 3, comprising the steps consisting in: calculating, according tosaid estimations, the control voltage to be applied to the motor, andderiving the triggering angle of the triac from this calculated voltage.20. A method for the digital control of a universal motor according toclaim 5, comprising the steps consisting in: calculating, according tosaid estimations, the control voltage to be applied to the motor, andderiving the triggering angle of the triac from this calculated voltage.21. A method for the digital control of a universal motor according toclaim 8, comprising the steps consisting in: calculating, according tosaid estimations, the control voltage to be applied to the motor, andderiving the triggering angle of the triac from this calculated voltage.22. A method for the digital control of a universal motor according toclaim 12, comprising the steps consisting in: calculating, according tosaid estimations, the control voltage to be applied to the motor, andderiving the triggering angle of the triac from this calculated voltage.